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TGI

1. Introduction

The QUIJOTE (Q-U-I JOint Tenerife Experiment) project, Refs. AYA2007-68058-C03 and AYA2010-21766-C03 within the Plan Nacional de Investigación, Desarrollo e Innovación program framework, has the main engineering goals of developing two instruments, the first one a Multi-Frequency Instrument (MFI) covering the 10 – 14 GHz and 16 – 20 GHz bands, and the second one, the Thirty-Gigahertz Instrument (TGI), a multi pixel receiver covering the range from 26 to 36 GHz. These two instruments should be exchangeable in a 3-m telescope suitably designed within the project. The information contained hereinafter is related with the TGI.

The TGI is accommodated in the minimum of the sky contribution in the microwave frequency range (see Fig. 1), between the water and the oxygen atmospheric absorption bands, in order to maximize the receiver sensitivity. Since the TGI is a multi-pixel camera with more than thirty receivers working together, this instrument increments significantly the sensitivity to detect the r parameter (tensor-to-scalar ratio) regarding the MFI.

Figure 1. Sky temperature expected at the observatory location (Izaña, Tenerife, Spain) with the bands of the different instruments defined with colored bars.

 

The QUIJOTE experiment operates at Izaña observatory, Tenerife, Spain, (see Fig. 2) with the telescope and the MFI taking measurements since 2012. It is expected that the TGI will be completely assembled and installed in the observatory for the commissioning phase by mid-2015; therefore, the TGI will start to gather meaningful data by the end of the year.

 

Figure 2. Projects dome with the two telescopes (left) and the MFI operating at Izaña observatory (right).

 

2. Principles of Operation

The main scientific goal of the QUIJOTE project is to combine the data from the ESA Planck mission and the MFI and TGI in order to study the physics of the inflationary period of the universe. A special emphasis is put on the detection of the primordial Gravitational Wave Background (GWB) with the goal of reducing the uncertainty of the r parameter in about an order of magnitude.

These scientific goals demand the design and operation of very high sensitive polarimeters (receivers capable of measuring the polarization state of the incoming electromagnetic signal). Therefore, suitable polarimeter schemes were designed for the MFI and the TGI. In the case of the TGI, the receiver scheme (hereinafter called pixel) is shown in Fig. 3.

 

Figure 3. Pixel scheme of the TGI.

 

The polarimeter obtains the signal polarization through the measurement of the so-called Stokes parameters I, Q and U simultaneously. The parameter V = 0 since it is assumed that the microwave background is linearly polarized. The linearly polarized incoming signal passes through the pixel feedhorn and reaches the polarizer and the orthomode transducer (OMT). At the OMT outputs there are two noise-like orthogonal signals proportional to the right-hand and left-hand components of the incoming signal, Er and El respectively. These two signals are amplified in the cryogenic low-noise amplifiers in the Front-End Module (FEM). These amplifiers are the key components of the receiver in order to determine the low noise performance and therefore to obtain a very sensitive instrument.

In the Back-End Module (BEM), which operates at room temperature, the signals are further amplified and filtered to define the operational bandwidth. A phase-switch module before the detection stage introduces different relative phase differences between pixel branches which help to minimize systematic errors in the receiver. Finally, the signals reach the detection module where they are correlated in two hybrid couplers and detected in Schottky diode detectors before being amplified with video amplifiers to accommodate the signals levels to the Data Acquisition System (DAS).

According with the scheme of Fig. 3, and assuming that the relative phase between pixel branches is zero, then

These signals, which are easy to obtain with simple mathematical calculations in the DAS, are proportional to the Stokes parameters, defined in a circular reference system as

Therefore, it is clear that

When the phase switches change their states in the module sixteen different states appear, four different relative phase differences repeated four times each. These changes produce that the Stokes parameters are obtained from the combination of different outputs through the whole cycle of the sixteen states, adding redundancy and therefore making the pixel less sensitive to systematic errors.

 

3. Number of Pixels

As stated before, the achievement of the scientific goals requires the use of very high sensitive receivers. In order to improve the instrument sensitivity even more, the number of pixels needs to be increased, since the number of pixels, N, and the instrument sensitivity are closely related as shown in (11), the radiometer’s equation.

Where K is a constant of proportionality which depends on the receiver configuration, Tsys is the system noise temperature, B is the pixel bandwidth, and t is the integration time, which is the time that the receiver is taking measurements.

From (11), it is clear that the larger the number of pixels is the smaller the temperature difference can be measured, that is, the higher the sensitivity is. Following this, the TGI instrument is equipped with 31 pixels, which are the maximum number of pixels that can be accommodated in the cryostat taking into account the focal plane maximum diameter defined in the project design and the footprint of each pixel.

 

4. Pixel Subsystems

4.1. Waveguide components before the Front-End Module

4.1.1. Feedhorn

The first component of the pixel, according with Fig. 3, is the feedhorn. It is a corrugated horn antenna designed to have more than 20 dBi of gain, very low cross-polarization values, more than 20 dB of return losses at the input port, and a Guassian profile with low side-lobe levels.

Figure 4. Artist view of the TGI feedhorn cross-section with main dimensions.

 

   

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Figure 5. Simulation and measurement comparison of the matching, directivity and cross-polarization (a) and measured radiation pattern at 32 GHz (b) of the feedhorn prototype.

 

4.1.2. Polarizer

The input signal at the feedhorn passes through a polarizer, which is a section of square waveguide provided with suitable designed ridges in its internal walls in such a way that the orthogonal components of the signal are 90° out of phase at the polarizer output. If the polarizer is placed within the pixel with its reference axis rotated 45° regarding the OMT reference system, then the orthogonal components of the input signal at the feedhorn output are converted in the right-hand and left-hand circular components at the polarizer output.

 

Figure 6. Artist view of the TGI polarizer cross-section with main dimensions.

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Figure 7. Measured performance of the 34 manufactured polarizer units: phase difference between orthogonal signals at the output (a); and port reflection following short ridge mode (b). Mean values are highlighted in red color.

 

4.1.3. Orthomode Transducer (OMT)

An orthomode transducer with in-phase outputs separates the left-hand and right-hand circular components of the incoming signal providing suitable signals to the receiver for the calculation of the Stokes parameters.

Figure 8. Artist view of the OMT internal configuration.

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Figure 9. Measured performance of one manufactured OMT: rectangular port reflection (a); and isolation (b). Mean values are highlighted in red color.

 

The previous subsystems are connected using suitable designed octagonal-shaped square-to-circular waveguide transitions in a very compact way. Fig. 10 shows the waveguide components manufactured for the TGI.

 

Figure 10. Waveguide components ready to be assembled in the pixels; the picture shows 34 polarizers (left-down corner), 34 OMT (right-down corner) and 27 feedhorn throats (up).

 

4.2. Cryogenic Low-Noise Amplifiers (Cryo-LNA)

A key component in the instrument performance is the cryo-LNA. It has to provide very high gain to minimize the noise contribution of the subsequent components while its own noise has to be kept as low as possible. For this reason, these amplifiers are cooled to cryogenic temperatures, around 20 K (-253 °C).

The cryo-LNAs designed for the TGI are assembled with two MMIC LNAs: the first one is designed using the 100 nm mHEMT technology from the IAF (Fraunhofer Institute, Freiburg, Germany); whereas the second one is designed with the 130 nm mHEMT technology from OMMIC. Between both MMICs there is a 5-dB attenuator that helps to improve the LNA stability. These devices are placed in a gold-plated aluminum chassis provided with WR-28 waveguide ports.

 

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Figure 11. Pictures of the cryo-LNA manufactured in series: full assembly (a); close view of the RF cavity (b).

  

     

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Figure 12. Performance of the 62 cryo-LNA measured at room temperature, 298 K, (a); and at cryogenic temperature, 12 K, (b). Mean values are highlighted in red and blue colors.

 

4.3. Gain and Filtering Modules

The first subsystem in the BEM is a module that provides further amplification and filters the signal to define the pixel effective bandwidth. The extra gain is obtained cascading two MMIC LNAs model AMMC6241 from OMMIC with a 10-dB attenuator in between to accomplish the gain requirements. The filter has been designed in microstrip technology on Alumina substrate. The microstrip technology enables the band-pass definition with low sharpness, which helps to compensate the bandwidth limitations of other subsystems in order to maintain the required effective bandwidth.

 

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Figure 13. Pictures of the assembled gain and filtering modules: RF cavity showing MMIC LNAs, 10-dB attenuator and filter (a), and a picture of units #49 to #66 (b).

 

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Figure 14. Measured performance of the gain and filtering modules: measured transmission gain of 66 units (a); and input matching of 66 units (b). Mean values are highlighted in red color.

 

4.4. Phase Switches Module

This module is composed of two radiofrequency branches which use a 180° and a 90° phase switches assembled together in each one in order to provide four phase states per branch. The module is designed with WR-28 waveguide ports. The phase switches are based on coplanar waveguide, slotline and microstrip transmission lines, and are assembled with PIN diodes to switch their individual phase state. The assembly inside the chassis is shown in Fig. 15, in which general, backside and detailed views are presented. The Phase Switches Module has the functionality of providing the sixteen phase states by the combination of its two branches. Then, the module is provided with TTL drivers, model DR65-0109 from MACOM Technology Solutions adding the full-switching capability to the module. The results of the module are shown in Fig. 16.

 

 

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Figure 15. Photographs of the TGI Phase Switches Module. (a) General view. (b) Backside of the chassis with the driver board. (c) Detailed view of the 90° (left part) and 180° (right part) phase switches.

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Figure 16. Phase Switches Module results. (a) Phase difference between states for each branch. (b) Transmission coefficient for state and branch.

 

4.5. Correlation and Detection Module

This module provides the output detected voltages (Vd1 to Vd4) correlating the input signals, proportional to Er and El, and detecting those using Schottky diode detectors. The module combines waveguide and microstrip technologies to accomplish all the functionalities.

The electrical scheme of the module was shown in Fig. 3. The input signals are divided using an in-phase waveguide power splitter. Then the divided signals are correlated in two microstrip 180° hybrid couplers designed on Alumina substrate. A waveguide 90° phase shifter in one branch enables to obtain the signals that are required to calculate the Stokes parameters according with Section 2.

 

 

Figure 17. Picture of a correlation and detection module.

 

The module contains four microwave detectors based on HSCH-9161 Schottky diode from Agilent Technologies developed in microstrip technology on Alumina substrate. They convert the radiofrequency signals into DC measurable voltages. The detectors have been designed and manufactured using 20 Ohm/square resistive layer transmission lines as a solution to simultaneously provide a flat sensitivity response and good return loss over the operating bandwidth of the receiver. A single unit of the detector and the sensitivity response in the frequency range from 24 to 38 GHz are shown in Fig. 18.

 

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Figure 18. Picture of the Schottky diode detector prototype (a); and measured sensitivity performance with an input power of -31 dBm (b).

 

After the detectors, the detected signals are amplified using video amplifiers in a differential configuration. Each video amplifier follows the electrical scheme presented in Fig. 18. A variable resistor (potentiometer) within the circuit enables to vary the gain of the video amplifier, which helps to accommodate the signal level to the DAS requirements.



Figure 19. Electrical scheme of the video amplifier for each output.

 

Different tests have been carried out to the correlation and detection modules. A frequency sweep with in-phase inputs is shown in Fig. 18a. As expected there is a maximum value in Vd1 and a minimum value in Vd2 across the band, which meets expressions in (1)-(4). The other two detected voltages are equal. Also, a test versus time is carried out with in-phase noise-like input signals and the results are presented in Fig. 18b. In this case, there are not ripples since the value is a mean detected value across the band. The different values in both plots in Fig. 18 are due to different setup conditions, mainly input powers.

 

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Figure 20. Measured detected voltages at the correlation and detection module outputs for module #2 (a); measured output voltages versus time for detection module #28 (b).

 

5. Power Budget and Output Voltages of the Radiometer

Following, the signals power levels and subsystems contributions are used to calculate the expected voltages at the pixel output working in a real environment in order to check the receiver configuration and its suitability to obtain adequate levels that can be used to extract the Stokes parameters; that is, the power budget. This calculation is based on realistic expected input signals and measured subsystems performance.

The input power to the FEM is calculated as:

                                                                                        

Where k is the Boltzmann’s constant (1.38·10-23 J/K), Tsys is the system noise temperature which includes all the contributions, and Beff is the effective bandwidth, which has been calculated with the available data so far, obtaining around 11.4 GHz.

The calculation of Tsys requires considering different contributions: the sky temperature, the antenna spillover, the noise of the waveguide components due to their losses, the cryo-LNAs noise temperature, and the BEM contribution which is greatly minimized by the cryo-LNAs gain.

According with the plot in Fig. 1, which was made with data taken at Izaña observatory, the sky temperature at the center frequency is around Tsky = 8 K. The antenna spillover has been estimated in 5 K; the contribution of all the waveguide components cooled at cryogenic temperatures has been calculated to be around 3 K; the cryo-LNAs are measured finding a mean noise temperature of 25 K and a gain higher than 42 dB which makes the contribution of the subsequent components negligible (see Fig. 12b). Therefore, the Tsys is around 41 K.

The obtained input power is around Pin = -82 dBm. This power is amplified by the FEM gain and the BEM gain, whereas the coaxial cables of the FEM-BEM connection and the phase switches module introduce noticeable losses resulting in a power level around -23 dBm at the correlation and detection module input. Considering the module losses, the detector sensitivity (around 1200 mV/mW), and the gain of the video amplifier, the expected voltage at the module output, ready to be digitalized in the DAS, is around 1.1 V (this voltage corresponds to the output with maximum value).